Detection

FIELD OF THE INVENTION

The invention relates to an interferometric sensor using a modulated phase detection scheme.

BACKGROUND OF THE INVENTION US 3 707 329 describes an apparatus for ana ^¬ lyzing a light beam modulated by a quantity to be measured. It includes a mechanical light chopper that modulates the amplitude of a light beam before entering an electro-opti ^¬ cal sensor.

DE 195 44 778 describes a current sensor where two sections of a sensing fiber of differing lengths are would around a conductor.

Sensors which rely on the interference between two waves, typically two orthogonal polarization modes of a wave, are known and used in a wide range of technical fields (see references [1]- [7]). The detector signals of these sensors are related to the cosine of the relative phase shift φ between the two waves. The phase shift can be detected using for example a polarimetric scheme. Such a sensor generally requires multiple detector channels (for example two quadrature polarimetric channels and a refer ^¬ ence channel measuring the total optical power) . In order to satisfy the high precision requirement in some applica ^¬ tions, e.g. DC electro-optic voltage sensing, a very good relative stability between multiple channels (-1-5% rela ^¬ tive channel stability for protection accuracy) has to be maintained, which is a serious practical challenge.

Interferometric sensors can also be built using the modulation phase detection (MPD) technique as described for example in reference [8], both in an open-loop and in a closed-loop configuration. The MPD technique is gener ^¬ ally implemented in a "non-reciprocal phase modulation" scheme and commonly used in fiber-optic gyroscopes (FOG) and fiber-optic current sensors (FOCS) [9, 10] . The co- owned Patent US7911196 [11] describes a voltage sensor in a reflective configuration incorporating a voltage sensing element (or several such elements), a 45° Faraday rotator, and the MPD modulation and detection electronics. A similar system with a transverse-configuration voltage cell can be found in [12] .

Whereas polarimetric optical phase measure ^¬ ments usually require the comparison of optical powers measured from multiple detectors, in MPD, a fast phase modulation is added onto the phase shift to be measured, and only one detector is needed to measure the response waveform and calculate the phase shift from it. The MPD phase shift calculation is independent of the absolute level of the measured power, and hence inherently immune to optical power fluctuations arising from low-frequency vibrations and environmental perturbations, which may vary between different detectors but cannot change the shape of the waveform a detector measures (provided that the per ^¬ turbation is much slower than the modulation frequency) .

Up to now, the design of a practical MPD sensor employs the "non-reciprocal phase modulation" principle. Such sensors can be largely divided into two classes: the Sagnac interferometer configuration and the reflective configuration. The optical design is reciprocal, meaning that all intrinsic phase shifts accumulated during propa ^¬ gation in the circuit must cancel out. The measurand- induced phase shift in the sensing element and the phase modulation imposed by the modulator, however, are non-reciprocal and add up. To this end, the interfering waves must pass twice in opposite directions through the phase modulator and any interconnecting fibers before reaching the detector. Single-mode (polarization-maintaining (PM) ) fibers are needed to transmit the waves from and to the optical phase modulator with a defined phase shift (and polarization) . For signal processing, both closed-loop and open-loop schemes have been developed to extract the measurand-induced phase shift from the measured waveform, see for example [8] .

The principle of reciprocal design has been essential in all practical MPD sensors, because the in ^¬ trinsic phase shifts in both the phase modulator and the PM fibers are highly sensitive to temperature or stress perturbations. Therefore, without the reciprocal optical circuitry, a simple MPD apparatus would not work reliably in a real-world environment, because the measured phase shift would be continuously perturbed by environmental dis ^¬ turbances. With the reciprocal design, on the other hand, a MPD sensor has been demonstrated to achieve a superb phase-measurement precision with a remarkable DC stabil ^¬ ity.

Implementing the reciprocal optical design for an all-optical-fiber device, such as a FOG or a FOCS, is relatively easy, because in such devices, an optical fiber is itself the sensing medium, and high coupling efficiency between the various fiber-optic components can be reliably achieved by standard splicing. However, for sensors in which the sensing element is not a fiber, but for example a bulk element, the reciprocal optical design with a fiber- coupled optical phase modulator is considerably harder to realize. That is because, after passing through the bulk- optic sensing element, the light must be coupled back into a single-mode fiber (in the reflective configuration the same fiber before the sensing element) , and to do that with high efficiency and reliability is by itself a substantial technical challenge.

In the light of the above, it is seen as an object of the invention to provide interferometric sensors using a differential modulation phase detection scheme, without the need of coupling light from the sensing ele- ment, which can be in particular a bulk-optic sensing element, into a single-mode fiber or waveguide.

SUMMARY OF THE INVENTION

Hence, according to a first aspect of the in ^¬ vention, an interferometric sensor is provided, with a sensing element whereby a measurand induces a relative phase shift between two waves, a phase modulator adding a modulation to the relative phase shift, at least two de ^¬ tectors, with a first detector detecting an interference signal responsive to a relative phase shift not including the measurand-induced relative phase shift, and with a second detector detecting an interference signal respon- sive to a relative phase shift including the measurand- induced relative phase shift. The two signals can be com ^¬ pared to derive a measurand value therefrom.

The term "wave" here is meant in the general physical sense of the word, including all types of oscil- lations traveling in space and time. The wave may have narrow or broad spectral content, may be long-lasting or be limited in duration, and may be generated by one source or be synthesized from multiple sources. The nature of the wave may in principle be mechanical (acoustic) , elec- tro-magnetic (optical), or be of any other type. In the following description, the invention is described using light waves as examples. The two interfering waves can be for example two orthogonal linear or circular polarization modes of a light wave.

In a preferred embodiment of this aspect of the invention, after being modulated by the phase modulator, the light waves pass through a beamsplitter before entering the sensing element, and two sets of polarizer and detector each measure an interference signal before and after the sensing element, respectively. Phase shifts are retrieved independently from the two measured waveforms, and their difference yields the phase shift inside the sensing me ^¬ dium, which is then converted to a measurand value.

Hence, applying the present invention, a sensor can be built with diminished sensitivity to phase shift changes in the wave path outside of the sensing element, while not requiring the waves to return along the same path to the phase modulator.

To maintain reasonably high coherence between the interfering waves at the detectors, preferably a group delay bias element can be introduced in the wave path to at least partially compensate the intrinsic relative group delays between the two waves, both before and after the sensing element.

The group delay compensation can be further enhanced by determining from the interference signals a value of the interference contrast or any related or equiv ^¬ alent measure. The interference contrast can be addition ^¬ ally used either to provide period information, which helps remove period ambiguity from the measured phase shifts and thus extend the unambiguous measurement range, or as a monitor signal for a controlled environment enclosing at least some of the birefringent elements of the sensor.

Advantageously, the modulation induced by the phase modulator is independent of the measurand. In addi ^¬ tion, the signal processing unit is adapted to determine the relative phase shift between the two waves from the modulation induced by the phase modulator. In particular, the modulation has at least one spectral component at a given frequency that is not present in the measurand, and the signal processing unit is adapted to determine the relative phase shift using the given frequency.

For voltage or electrical field measurements according to this invention, the sensing element can comprise an electro-optic crystal, a crystalline electro-op ^¬ tic fiber, a poled fiber, or a fiber or bulk optic material attached to a piezo-electric element. For force or strain measurements according to this invention, the sensing el- ement can comprise an optical fiber or a bulk optic mate ^¬ rial. For optical magnetic field sensors or current sen ^¬ sors according to this invention, the sensing element can comprise optical fibers or waveguides, including specialty low birefringent fibers, flint glass fibers, or spun highly-birefringent fibers, bulk magneto-optic materials, such as yttrium iron garnet crystals or fused silica glass blocks, or optical fibers, waveguides, or bulk optical ma ^¬ terials attached to a magneto-strictive element or combi ^¬ nations thereof.

The sensor is particularly preferred as a sensor for DC signals and more particularly for DC voltage or electrical field measurements, especially for medium or high voltage applications. It can however also be poten ^¬ tially applied to a fiber-optic current sensor, a rotation sensor, or other MPD sensors.

It is particularly suited for a sensor having a bulk sensing element.

The above and other aspects of the present in ^¬ vention together with further advantageous embodiments and applications of the invention are described in further de ^¬ tails in the following description and figures.

BRIEF DESCRIPTION OF THE FIGURES

FIG. 1 illustrates a first example in accord ^¬ ance with the present invention; FIGs. 2A and 2B show examples of the invention using relative group delay compensation; and FIG. 3 illustrates an example of the invention in a reflective configuration.

DETAILED DESCRIPTION

A basic differential MPD sensor example ac ^¬ cording to the invention is shown in FIG. 1. A light source 1 generates a light wave, which is guided via a single-mode fiber to a fiber optic polarizer 2. The po ^¬ larized light output, through a 45° fiber splice 4 and carried in both axes of PM fibers 3, passes through an optical phase modulator 5 to a fiber-optic collimator 6, where the light is coupled out of the PM fiber 3. A birefringence-free beamsplitter 7 is located between the collimator and a sensing element 8. One branch of the light output split by the beamsplitter 7 passes through a first bulk linear polarizer 9-1, and is detected by a first photodetector 10-1. The other branch of the light output passes through the sensing element 8, a second bulk linear polarizer 9-2, and is detected by a second photodetector 10-2. The output signals of the two detectors 10-1, 10-2 are inputs to a signal processing unit 11. The optical axes of the sensing medium 8 are aligned parallel, and the axes of the bulk linear polarizers 9-1, 9-2 are aligned at 45° to the axes of the PM fiber 3 connected to the phase modulator 5. The optical phase modulator may be an inte- grated-optic LiNb03 birefringent phase modulator [8, 10], or a Y-type modulator with a 90° PM fiber splice in one of the branches as described in reference [9] . Another option is a phase modulator consisting of a piezoelectric trans ^¬ ducer with the fiber attached to it [9] . The phase modulator 5 adds a phase modulation onto the relative phase shift between the two interfering waves, and the signal processing unit performs independent MPD signal processing using the waveforms measured at the two photodetectors 10-1 and 10-2, and calculates two phase shifts respectively. The waveform measured at detector 10-1 yields a first phase shift φι between the splice 4 and the polarizer 9-1, and the waveform measured at detector 10-2 yields a second phase shift φ2 between the splice 4 and the polarizer 9-2.

The difference of the two measured phase shifts Δφ = φ2 - φι therefore corresponds to the phase shift inside the sensing medium 8. The differential phase measurement Δφ is not affected by any phase shift changes between the polarizer 2 and the beamsplitter 7, provided that the MPD modulation frequency is much higher than the perturbation frequency, so that the perturbation is simultaneously and equally tracked by the φι and φ2 measurements and is there ^¬ fore fully subtracted.

For voltage measurements, an electro-optic crystal can be used as the sensing medium 8 to convert the voltage to be measured to a phase shift between the or ^¬ thogonal polarization modes in the crystal. An electro- optic crystal without natural birefringence (such as Bi Ge30i2, BGO) is preferred. An electro-optic waveguide or fiber can also be used as the voltage sensing medium. The end faces of the BGO crystal (cut along the [001] direction) are electrically connected to the electrodes that provide the voltage drop V.

As mentioned, there are two types of MPD signal processing configurations (see also ref. [8]): open-loop and closed-loop. Generally speaking, the closed-loop con ^¬ figuration is more difficult to implement than the open- loop counterpart, but is superior in the stability and linearity of the sensor output.

In the open-loop configuration, the phase mod ^¬ ulator exerts a free-running modulation, and the phase shift is calculated from the measured waveform by the sig ^¬ nal processing unit 11. A typical implementation is described in Patent US6429939 (reference [13]), in which a sinusoidal phase modulation P(t) is used, and the phase shift is calculated from the relative ratios of the various harmonic powers in the measured waveform.

In an open-loop sensor, no feedback to the mod ^¬ ulation waveform is made from the calculated phase shift, except that the modulation amplitude may possibly (but not imperatively) be stabilized using an output calculated from the measured waveform.

In the closed-loop configuration [8], in contrast, the offset of the phase modulation waveform (or equivalently the phase of a 2n-amplitude sawtooth waveform) is constantly adjusted by the opposite of the calculated phase shift via a feedback loop, so that the phase shift working point of the sensor is always maintained at a fixed point (e.g. zero). A detector and the phase modulator are linked via the signal processing unit 11 in a feedback loop, with the retrieved phase shift fed back to control the phase modulation waveform.

In the differential MPD scheme of the present invention, there are at least two detectors but usually only one phase modulator. Therefore, it is not possible to run closed-loop detection on both detector outputs. One can either implement open-loop signal processing on both detector outputs; or alternatively, one can implement closed-loop signal processing on one of the detector out ^¬ puts (for example detector 10-1), and use open-loop signal processing to extract phase shift from the other detected signal (for example detector 10-2) .

If one chooses to implement closed-loop signal processing on detector 10-1, then the phase shift from the splice point 4 to the polarizer 9-1 will always be main- tained at a fixed set value (e.g. zero) . In this case, the phase shift in the sensing medium 8 is simply the phase shift measured at the detector 10-2 minus the set value, and no further subtraction between the detected signals is in principle needed.

A second aspect of the invention relates to the coherence of the light generated by the light source 1. Although the MPD phase measurement principle would work with a highly coherent light source such as a narrowband laser diode, in practice a broadband source with a short coherence time, such as a superluminescent light-emitting diode (SLED) , is often preferred, in order to localize interfering waves and avoid spurious interference from cross-coupling at various fiber junctions. With such a low-coherence light source, interference signals can only be observed in a relatively short range where the relative group delay between the two polarizations is well within the source coherence time, beyond which the interference contrast would drop quickly and eventually to zero, and the MPD phase detection scheme would cease to work.

Therefore, if a low-coherence light source is used, the various birefringent components in the system, such as the PM fibers and modulators, must be designed and adjusted to balance their intrinsic relative group delays, such that the total relative group delay between the splice point 4 and the output polarizers 9-1 and 9-2 is kept minimal and falls well within the coherence time of the light source. For this purpose, it can be preferable to insert an additional group delay compensation element into the wave path.

Nevertheless, as environmental conditions change, the temperature and stress sensitivities of the components and the possible inhomogeneous environment in the system can still cause the total relative group delay to slowly drift away from the minimal design value. There ^¬ fore, in a sensor system where the environmental disturb- ances are too large and/or the coherence time of the light source is too short, it may prove beneficial to actively stabilize the total relative group delay, or equivalently the interference contrast, in order to successfully imple ^¬ ment the differential MPD scheme. This of course requires the capability to measure and control the interference contrast of the MPD signal.

Although the MPD scheme has conventionally only been used to measure the phase shift, it is possible to extend the signal processing algorithm and also derive the interference contrast from the measured signal. Numerous implementations exist, depending on factors such as the modulation waveform and feedback control configuration. In the following, an example procedure for the interference contrast calculation is described with a sinusoidal modu ^¬ lation waveform in an open-loop configuration, which is an extension of the algorithm described in Patent US6429939 (reference [13]).

The phase modulator 5 is in series with the sensing medium 8, adding an additional phase shift modula ^¬ tion P(t) to the phase shift to be measured φ. A detector such as detector 10-2 in FIG. 1 measures the modulated optical power after a linear polarizer such as polarizer 9-2. The modulated detector signal can be written as

[1] I(t) = I ₀ /2 [1 + A cos^ + p(t))]. with Io representing the output power of the light source, φ being the phase shift to be measured at the center wave ^¬ length of the waves, and A being the interference contrast.

With a sinusoidal modulation P(t)= Psin(Qt), the detector signal of eq. [1] can be written in a Fourier expansion as a series of harmonics at different orders k of the modulation frequency Ω, i.e.,

[1' ] I (t) = ∑[B _k cos (kQt + ξ]<) ] with the first three harmonic amplitudes Bk and phases ξ¾ being Bo = (IO / 2) [1 + A Jo (β) cos (φ) ] , ξθ = 0 B ₂ = IO J ₂ (β) cos (φ) , ξ2 = 0 using Bessel functions of the first kind Jk(P) · The signs of the harmonic components can be ascertained by comparing the phases of the harmonic components with that of the excitation waveform.

A vector or complex number can be formed from the above representation, which allows to derive the phase shift principal value cp and the interference contrast A from the detected signal.

[2] Y = B ₂ / J2(P) + i Bi / Ji(p)= Io A exp(i φ)

The phase shift principal value cp can be cal ^¬ culated as the argument of Y, and the interference contrast A equals its absolute value divided by Io. The total optical power Io can be calculated as

IO = 2Bo - Jo(p) abs(Y) cos(arg(Y)).

A preferred modulation amplitude is P = 1.84 rad where Ji (β) has its first maximum; another preferred amplitude is P = 2.63 rad where Ji (β) and J2 (β) are equal. However, in principle for the MPD method to work, the phase modulation amplitude P can be arbitrarily small. Furthermore, it is known that the modulation amplitude P can also be calculated from the measured harmonic amplitudes, e.g., for the purpose of stabilizing the amplitude. Further details of the open-loop MPD signal processing can be found in the references [8, 13] .

In the example of FIG. 1, the additional meas ^¬ urement of the interference contrast or any parameter re- lated or equivalent to it is denoted as IC.

Meanwhile, since the interference contrast of the MPD signal can be disturbed by the environment, it can also be controlled by controlling an environmental condi ^¬ tion. As indicated by the arrow IC in FIG. 1, the deter ^¬ mined value of the interference contrast or any related signal can be used to control at least some parameters of the sensor.

One possibility is to control the temperature of the phase modulator in a heating device or a Peltier cell. As the temperature changes, the birefringence and the relative group delay of the modulator crystal also change. Similarly, one can also control the temperature or strain of a section of a PM fiber. Other methods of implementing such an interference contrast control can also be readily devised.

Employing both the measurement and the control mechanisms, a feedback loop can then be established to stabilize the MPD interference contrast in order to prevent it from drifting out of the coherence range. The feedback, however, need neither be accurate nor be fast, as the co ^¬ herence range of the light source is generally moderate, and the possible group delay drift is also slow in nature.

Next, a few examples of relative group delay compensation are presented in the design of a sensor.

In the example of FIG. 2A, two essentially identical birefringent phase modulators 5 and 5' are spliced together with a 90° relative axis alignment, so their intrinsic birefringence cancels. The length of PM fiber 3 between the 45° splice point 4 and the 90° splice point 4' should be the same as the PM fiber length between the 90° splice point 4' and the collimator 6, so that their birefringence also cancels. The phase modulator 5' and the PM fiber between splice point 4' and collimator 6 together constitute a group delay compensation element 20. At least some of the birefringent components can be kept in an enclosure 21 to maintain a uniform environment among them and to shield against stress and temperature fluctu ^¬ ations. The PM fiber section 3 left outside the enclosure 21 is preferably kept as short as possible. Meanwhile, the MPD phase modulation can be distributed between two phase modulators 5 and 5' , allowing each modulator to op ^¬ erate in a smaller range, possibly with a better linearity.

The example of FIG. 2B is similar to FIG. 2A, with the difference being that only one phase modulator 5 is used, and the intrinsic birefringence of the phase mod ^¬ ulator crystal 5 is compensated by an extra length in one of the sections of the PM fiber 3 (indicated in FIG. 2B by loops) acting as a group delay compensation element 20. An enclosure 21 can likewise contain at least some of the birefringent components. Because the temperature charac ^¬ teristics of the phase modulator crystal 5 and the PM fiber 3 can be dissimilar, this design may suffer more group delay drift than the design shown in FIG. 2A, as the tem- perature of the sensor changes.

The measurement of IC has a further benefit with respect to phase ambiguity removal and measurement range extension, if a low coherence light source is used as source 1, as can be explained as follows:

The phase shift measurement of an interference sensor is generally intrinsically 2n-periodwise ambiguous. However, the autocorrelation function (herein determined as the interference contrast) of a low-coherence source is a narrow function, whose value changes significantly from one phase period to another within its coherence time. Therefore, provided that the phase shift principal value (phase shift mod 2n within (-n, +n] ) and the interference contrast are simultaneously measured, in a range where the autocorrelation function has a strong monotonic dependence on the relative group delay, the interference contrast can be used to allocate the measured phase shift principal value to the correct period, and thereby unambiguously de ^¬ termine the full value of the relative phase shift.

Hence the simultaneous measurement of the rel- ative phase shift and the IC (or any related parameter) , as per eq. [2] for example, can be used in a sensor of the present invention to extend the phase shift measurement range beyond 2n, effectively to a range determined by the slope and form of the auto-correlation function of the light source (or of the cross-correlation function, if the waves are generated by two different sources) .

Any other period disambiguation method, such as using two different wavelengths, can also be used in combination with the present invention, in order to extend the unambiguous phase shift measurement range beyond 2n. Examples of such methods are described in the patent ap- plications WO9805975A1 [6] and EP1179735A1 [7] . A third aspect of the present invention relates to the geometric design of the sensor. FIG. 1 shows an example of a trans- missive-configuration sensor, in which the light source 1 (and many other components) and detector 10-2 are located at two opposite ends of the sensing crystal. For the example of a voltage sensor, this results in several com ^¬ ponents, especially one of the detectors 10-1 or 10-2, to be located at a high voltage potential. FIG. 3 illustrates an example of a reflective-configuration voltage sensor. In the reflective configuration, a reflecting optic 30 is placed at one end (preferably the end at the high-voltage potential) of the sensing element 8, while all the other optics are located at the other end (preferably the end at the ground potential) . The reflecting optic 30 may be a flat/curved mirror, a roof mirror, a corner-cube retrore- flector, or simply a reflective thin film coating deposited on the end face of the crystal. The reflection at the reflecting optic 30 should ideally preserve the polariza ^¬ tion state of the light without rotation or polarization- dependent phase shift. The element 20 denotes again an optional group delay compensation element or elements as described when making reference to FIGs. 2A and 2B above.

The electro-optic axis of the crystal should be aligned parallel to the axes of the PM fiber 3. Pref- erably, the beamsplitter 7 before the sensing medium 8 should be aligned with its axes at 45° relative to the PM fiber axes, in order to equalize any possible phase shift the two interfering waves may experience from the beamsplitter. Any residual system phase shifts, for example from the beamsplitter, the retroreflector or from the residual natural birefringence of the BGO crystal, can be characterized and taken out by calibration. The residual birefringence of BGO can also be reduced by combining two BGO crystals in series, with antiparallel [001] axes and the x/y axes rotated 90° against each other. In this arrangement, the electro-optic phase shifts add up, while the intrinsic birefringence cancels, leading to a better zero-point stability. The detectors 10-1 or 10-2 may be directly attached to the remaining parts of the sensor, or, alternatively, they may be connected to the sensor via single-mode or multimode optical fibers.

While some preferred embodiments of the inven ^¬ tion are shown and described herein, it is to be understood that the invention is not limited thereto but may be oth ^¬ erwise variously embodied and practiced within the scope of the following claims.

REFERENCES CITED

[1]G. A. Massey, D. C. Erickson, and R. A.

Kadlec, "Electromagnetic field components: their measure ^¬ ment using linear electrooptic and magnetooptic effects, " Appl. Opt., vol. 14, pp. 2712-2719, 1975.

[2]K. Bohnert and J. Nehring, "Method and de- vice for the optical determination of a physical quan ^¬ tity, " US5715058, 1998.

[3]R. C. Miller, "Electro-optical voltage measuring system incorporating a method and apparatus to derive the measured voltage waveform from two phase shifted electrical signals," US Patent US4904931, 1990.

[4]R. C. Miller, "Method of deriving an AC waveform from two phase shifted electrical signals, " US Patent US5001419, 1991.

[5]K. Kurosawa, S. Yoshida, E. Mori, G.

Takahashi, and S. Saito, "Development of an optical in ^¬ strument transformer for DC voltage measurement, " Power Delivery, IEEE Transactions on, vol. 8, pp. 1721-1726, 1993.

[6]0. Beierl, T. Bosselmann, and M. Willsch, "Method and arrangement for optically detecting an elec ^¬ trical variable," WO9805975A1, 1998.

[7]M. Stanimirov, U. Meier, K. Bohnert, and J. Glock, "Method of measuring a voltage and voltage sen ^¬ sor," EP1179735A1, 2002.

[8]H. Lefevre, The Fiber-Optic Gyroscope: Ar- tech House, 1993.

[9]K. Bohnert, P. Gabus, J. Nehring, and H. Brandle, "Temperature and vibration insensitive fiber-op ^¬ tic current sensor, " Journal of Lightwave Technology, vol. 20, pp. 267-276, 2002.

[10] K. Bohnert, P. Gabus, J. Nehring, H.

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LIST OF REFERENCE SIGNS

light source 1

fiber-optic linear polarizer 2

polarization maintaining (PM) fiber 3 fiber splice 4, 4' optic phase modulator 5, 5' collimator 6

beamsplitter 7

sensing element 8

linear polarizer 9-1,9-2

optical detector 10-1,10-2

signal processing unit 11

group delay bias element 20

controlled enclosure 21

reflecting optic 30