TAN NIANXIONG (SE)
| WO1995006977A1 | 1995-03-09 |
| US5495201A | 1996-02-27 | |||
| US4839542A | 1989-06-13 | |||
| US4686487A | 1987-08-11 |
| 1. | A method for continuoustime filtering in digital CMOS process, c h a r a c t e r i z e d by using current mirrors to realize continuoustime filters in a digital CMOS process, where pole frequencies are determined by a transconductance of an MOS transistor and a capacitance of a capacitor seen by its gate, wherein the capacitance defining the pole frequency can take any form including an offchip capacitor. |
| 2. | A device for continuoustime filtering in a digital CMOS process", c h a r a c t e r i z e d in that current mirrors are used to realize continuoustime filters in a digital CMOS process, where pole frequencies are determined by a transconductance of an MOS transistor (6) and a capacitance of a capacitor (8) seen by its gate, wherein the capacitance defining the pole frequency can take any form including an offchip capacitor. |
| 3. | A device according to claim 2, c h a r a c t e r i z e d in that a current mirror consisting of transistors MO (6) , Ml (7) and a gate capacitor or offchip capacitor C0 (8) are used to determine the pole frequency. |
| 4. | A device according to claim 2, c h a r a c t e r i z e d in that two or more current mirrors are provided to be cascaded to realize higher order filters and in that ntypε ("1") and ptype ("2") current mirrors are provided to be alternate to save power dissipation. |
| 5. | A device according to claim 2, c h a r a c t e r i z e d in that two or more current mirrors are provided to be directly cascaded to realize higher order filters and to reduce the spread in pole frequencies by only using ntype or ptype current mirrors. |
FIELD OF THE INVENTION
The present invention relates to a method for continuous-time filtering in digital CMOS process and a device for continuous- time filtering in digital CMOS process.
BACKGROUND OF THE INVENTION
It is of importance to design a mixed analog/digital system in digital CMOS process concerning processing cost, testing cost and performance. There has been strong interest in designing sampled data systems, e.g. switched-current filters and data converter in digital CMOS processes, see for example C. Toumazou, J.B. Hughes, and N.C. Battersby (Eds) , "Switched- Currents: an Analogue Technique for Digital Technology,", Peter Peregrinus Ltd., 1993, and N. Tan, "Switched-current delta-sigma A/D converters", J. Analog Integrated Circuits and Signal Processing, Jan. 1996, pp 7-24. However, to utilize these kind of techniques, antialiasing filters are usually needed before sampling the analog input in order to avoid aliasing. Traditionally, a separate chip using analog CMOS process or discrete RC filter circuitry is used. Obviously, integrating the continuous-time filters, or antialiasing filters with sampled data systems and DSP circuits on the same chip offers the best performance/cost ratio.
In, for example, N. Tan and M. Gustavsson, Voltage-to-current converter", pending US patent application No. 08/646,964, May 8,
1996, a method was specifically developed to realize a low-pass filtering function embedded with a voltage-to-current conversion.
In, for example, US-A-4 , 839,542 there are disclosed active transconductance .filters, which belong to a filter type which is called a transconductance-capacitance (gm-C) filter. The basic idea is to create poles by using linear capacitors and transconductors. As for most active components, current mirrors are used as active loads for the transconductors and current mirrors are not utilized to create poles for any filtering purposes.
In O95/06977, current mirrors are disclosed and only used as active loads to increase the gain for the amplifier. As a matter of fact, for most gain stages, current mirrors are used as active loads to increase the gain.
In US-A-4, 686,487 there is disclosed how to design current mirrors for amplifiers in order to have high speed operation. The pole due to the current mirror is parasitic and the means of adding a resistor is invented to reduce the effect on high speed operation.
SUMMARY OF THE INVENTION
The invention relates preferably to the design of continuous- time filters for sampled data systems in digital CMOS processes. In a digital CMOS process neither resistors nor linear capacitors are available. Therefore it is not possible or simply not practical to design continuous-time filters using
traditional methods. It has been proposed to utilize current mirrors to realize filtering functions in a voltage-to-current converter. The pole frequency is therefore determined by the transconductance of an MOS transistor and the capacitance seen at its gate. In this application, a generalized method of designing continuous-time filters in digital CMOS process and methods of cascading have been proposed to reduce the spread of the pole frequencies.
BRIEF DESCRIPTION OF THE DRAWINGS
Fig. 1 " is a circuit showing a basic current mirror as a single- pole filter.
Fig. ' 2 is a graph showing SPICE simulation results of fig. 1, wherein cascode current mirrors and cascode current sources are used and the capacitor is realized by NMOS transistors.
Fig. 3 a and b are circuits showing cascading techniques according to the invention.
Fig. 4 is a graph showing SPICE simulation results of fig. 3b, wherein cascode current mirrors and cascode current sources are used and the capacitors are realized by NMOS transistors.
BRIEF DESCRIPTION OF THE PREFERRED EMBODIMENTS
In digital CMOS processes, neither resistors nor linear capacitors are available. Though it is possible to utilize the gate poly as resistors, the sheet resistance is very small and has large variation for a sub-micro CMOS process, and well resistors are sensitive to noise and have large variation as well. Therefore, active components are intended to be used, i.e. transistors, to realize resistance. Though it is possible to
utilize the single poly layer and metallizations to realize a linear capacitor, the sheet capacitance is very small in a sub- micron CMOS process. Therefore, the gate capacitance is intended to be utilized, which has much larger sheet capacitance. The basic current mirror used as a single-pole low pass filter is shown in Fig. 1.
The capacitor C 0 1 can be realized by a gate capacitor on chip, or realized by an off-chip capacitor, if the cut-off frequency of the filter is required to be very low. By properly dimensioning the sizes of transistors M 0 2 and M λ 3 and their associated bias currents 4, 5, a scaling factor can also be realized within this filter.
The pole frequency of the single-pole filter shown in fig. 1 is given by
where g m0 is the transconductance of the diode-connected transistor M 0 2 and C p0 represents all the parasitics at the gate of transistor M 0 2.
The nonlinearities in the transconductances do not introduce distortion in the output current as long as the transconductances of M 0 2 and M x 3 are matched or constantly rationed. However, nonlinearities in the capacitance can introduce error in the output current . Though the gate capacitance is highly nonlinear across the whole operation region, in a current mirror configuration as shown in fig. 1,
the gate voltage change is quite limited, making the transistors operate in a well specified region all the time. Therefore, the gate capacitance does not vary dramatically and the linearity is acceptable. When external capacitors are used, linearity can also be guaranteed.
However, the transconductance of a transistor is dependent of the drain current, i.e.,
where μ n is the channel charge mobility, C ox is the unit gate capacitance, W/L is the transistor size, and i D is the drain current. Therefore, when the drain current in transistor M 0 2 changes accommodating input current I 0 , the transconductance g ra0 changes, making the pole frequency change. In fig. 2 it is shown the SPICE simulation results, when the input current changes between ±0, 5 I bias0 .
It can be seen that the circuit of fig. 1 is a single-pole system, having 20 dB/dec frequency roll-off. And the change in the 3-dB frequency is well in line with the prediction given by the equation of transconductance. The change in the pole frequency also introduces distortion, when the input signal frequency approaches the cut-off frequency, in that a different input amplitude experiences a different attenuation.
The simulated total harmonic distortion is about -50 dB, when the input is a 100 Khz sinusoidal with amplitude equal to one- fourth of the bias current. When the input frequency decreases
to 10 Khz, the total harmonic distortion is less than -70 dB. When the input frequency is larger than the cut-off frequency, the total harmonic distortion is attenuated by the filter itself .
Obviously, to make the pole frequency well-defined, the change in the drain current is needed to be as small as possible. One way to do so is to limit the input current compared with the bias current. This is very power consuming. However, proper cascading realizing higher-order filters can reduce the variation in the pole frequencies.
To increase the filter order and reduce the variation in pole frequencies cascading of current mirrors can be used. A single- pole system only gives a 20-dB/dec roll-off. In many applications, sharper cut-off is needed. Cascading two single- pole systems realizes a two-pole system having a 40-dB/dec roll- off. Sharper cut-off can be realized by cascading more stages. There are two possibilities of cascading as shown in fig. 3a and b.
The use of cascading shown in fig. 3a results in lower power consumption due to the use of the p-type branch. The n-type branch "1" consists of n-type transistors M0 6 and Ml 7, capacitor C 0 8 and bias current IbiasO 9 for transistor M0 6. The p-type branch "2" consists of p-type transistors M2 10 and M3 11 capacitor C 1 12 and bias current Ibiasl 13 for M3 11. The n-type branch is similar to the one shown in fig. 1 except that the bias current for Ml 7 is omitted due to the use of the p- type branch. Transistors Ml 7 and M2 10 bias each other. The p- type branch is the same as the n-type except p-type transistors
are used. However, this kind of cascading influences the pole frequencies. Suppose that input current I 0 is positive, then the drain current in M 0 6 increases making its transconductance to increase. Therefore, the pole frequency determined by the transconductance of M 0 6 and capacitor C 0 8 will increase. At the same time, the drain current in M 2 10, equal to the drain current of M 7, increases as well making its transconductance to increase. Therefore, the pole frequency determined by the transconductance of M 2 10 and the capacitance of C λ 12 will increase as well . The combined effect is that the pole frequencies vary more rapidly as the input current varies.
The cascading technique shown in fig. 3 b results in more power consumption due to an extra n-type branch. It consists of two n- type branches "1" and "2" , which are exactly the same as the one shown in fig. 1. However, it has a big advantage stabilizing the pole frequencies. Suppose that input current I 0 is positive, then the drain current in M0 6 increases making its transconductance increase. Therefore, the pole frequency determined by g m0 /C 0 will increase. At the same time, the drain current in M 2 10 decreases making its transconductance decrease. Therefore, the pole frequency determined by will decrease. The combined effect is that the variations in the two pole frequencies tend to reduce the total variation.
In fig. 4 the SPICE simulation results are shown, when the input current changes between ±0,5 I b i s o-
It can be seen that the circuit of fig. 3b is a two-pole system, having 40-dB/dec frequency roll-off. And the change in the variation in the 3-dB frequency is reduced considerably.
The simulated total harmonic distortion is less than - 60 dB, when the input is a 100 Khz sinusoidal with amplitude equal to one-fourth of the bias current. When the input frequency decreases to 10 Khz, the total harmonic distortion is less than -80 dB. When the input frequency is larger than the cut-off frequency, the total harmonic distortion is attenuated by the filter itself.
While the foregoing description includes numerous details and specificities, it is to be understood that these are merely illustrative of the present invention, and are not to be construed as limitations. Many modifications will be readily apparent to those skilled in the art, which do not depart from the spirit and scope of the invention as defined by the appended claims and their legal equivalents.